SC4501EVB datasheet(10/21 Pages) SEMTECH

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POWER MANAGEMENT

SC4501

Application Information

The absolute maximum operating frequency of the

converter is therefore

MHz

150

MIN

. The

actual operating frequency needs to be lower to allow for

modulating headroom.

The power transistor in the SC4501 is turned off every

switching period for an interval determined by the

discharge time of the oscillator ramp and the propagation

delay of the power switch. This minimum off time limits

the maximum duty cycle of the regulator at a given

switching frequency. A boost converter with high

OUT

ratio

requires long switch on time and high duty cycle. If the

required duty cycle is higher than the attainable maximum,

then the converter will operate in dropout. (Dropout is a

condition in which the regulator cannot attain its set

output voltage below current limit.)

The minimum off times of closed-loop boost converters set

to various output voltages were measured by lowering their

input voltages until dropout occurs. It was found that the

minimum off time of the SC4501 ranged from 80 to 110ns

at room temperature.

Beware of dropout when operating at very low input voltages

(1.5-2V) and with off times approaching 110ns. Shorten

the PCB trace between the power source and the device

input pin, as line drop may be a significant percentage of

the input voltage. A regulator in dropout may appear as if

it is in current limit. The cycle-by-cycle current limit of the

SC4501 is duty-cycle and input voltage invariant and is

typically 2.8A. If the switch current limit is not at least 2A,

then the converter is likely in dropout. The switching

frequency should then be lowered to improve controllability.

Both the minimum on time and the minimum off time

reduce control range of the PWM regulator. Bench

measurement showed that reduced modulating range

started to be a problem at frequencies over 2MHz. Although

the oscillator is capable of running well above 2MHz,

controllability limits the maximum operating frequency.

Inductor Selection

The inductor ripple current

∆I

L of a boost converter

operating in continuous-conduction mode is

(

)

CESAT

−

∆

(5)

where f is the switching frequency and L is the inductance.

Substituting (3) into (5) and neglecting V

CESAT ,









−

∆

OUT

(6)

In current-mode control, the slope of the modulating

(sensed switch current) ramp should be steep enough to

lessen jittery tendency but not so steep that large flux swing

decreases efficiency. Inductor ripple current

∆I

L between

25-40% of the peak inductor current limit is a good

compromise. Inductors so chosen are optimized in size

and DCR. Setting

∆I

L = 0.3•(2) = 0.6A, VD=0.5V in (6),













−













−

∆

OUT

(7)

where L is in

µH and f is in MHz.

Equation (6) shows that for a given V

OUT, ∆IL is the highest

when

(

)

OUT

. If V

IN varies over a wide range, then

choose L based on the nominal input voltage.

The saturation current of the inductor should be 20-30%

higher than the peak current limit (2.8A). Low-cost powder

iron cores are not suitable for high-frequency switching

power supplies due to their high core losses. Inductors

with ferrite cores should be used.

Input Capacitor

The input current in a boost converter is the inductor

current, which is continuous with low RMS current ripples.

A 2.2-4.7µF ceramic input capacitor is adequate for most

applications.

Output Capacitor

Both ceramic and low ESR tantalum capacitors can be

used as output filtering capacitors. Multi-layer ceramic

capacitors, due to their extremely low ESR (<5m

Ω), are

the best choice. Use ceramic capacitors with stable

temperature and voltage characteristics. One may be

tempted to use Z5U and Y5V ceramic capacitors for

output filtering because of their high capacitance and

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