OPA842IDBVT datasheet(14/19 Pages) TI

14 / 19 page

OPA842

SBOS267A

www.ti.com

21MHz bandwidth shown in the Electrical Characteristics

agrees with that predicted using the simple formula and the

typical GBP of 200MHz.

OUTPUT DRIVE CAPABILITY

The OPA842 has been optimized to drive the demanding load

of a doubly-terminated transmission line. When a 50

Ω line is

driven, a series 50

Ω into the cable and a terminating 50Ω load

at the end of the cable are used. Under these conditions, the

cable’s impedance will appear resistive over a wide frequency

range, and the total effective load on the OPA842 is 100

Ω in

parallel with the resistance of the feedback network. The

electrical characteristics show a +2.8V/–3.3V swing into this

load—which will then be reduced to a +1.4V/–1.65V swing at

the termination resistor. The

±90mA output drive over tem-

perature provides adequate current drive margin for this load.

Higher voltage swings (and lower distortion) are achievable

when driving higher impedance loads.

A single video load typically appears as a 150

Ω load (using

standard 75

Ω cables) to the driving amplifier. The OPA842

provides adequate voltage and current drive to support up to

three parallel video loads (50

Ω total load) for an NTSC

signal. With only one load, the OPA842 achieves an excep-

tionally low 0.003%/0.008

° dG/dP error.

DRIVING CAPACITIVE LOADS

One of the most demanding, and yet very common, load

conditions for an op amp is capacitive loading. A high-speed,

high open-loop gain amplifier like the OPA842 can be very

susceptible to decreased stability and closed-loop response

peaking when a capacitive load is placed directly on the

output pin. In simple terms, the capacitive load reacts with

the open-loop output resistance of the amplifier to introduce

an additional pole into the loop and thereby decrease the

phase margin. This issue has become a popular topic of

application notes and articles, and several external solutions

to this problem have been suggested. When the primary

considerations are frequency response flatness, pulse re-

sponse fidelity, and/or distortion, the simplest and most

effective solution is to isolate the capacitive load from the

feedback loop by inserting a series isolation resistor between

the amplifier output and the capacitive load. This does not

eliminate the pole from the loop response, but rather shifts it

and adds a zero at a higher frequency. The additional zero

acts to cancel the phase lag from the capacitive load pole,

thus increasing the phase margin and improving stability.

The Typical Characteristics show the recommended “RS vs

Capacitive Load” and the resulting frequency response at the

load. The criterion for setting the recommended resistor is

maximum bandwidth, flat frequency response at the load.

Since there is now a passive low-pass filter between the

output pin and the load capacitance, the response at the

output pin itself is typically somewhat peaked, and becomes

flat after the roll-off action of the RC network. This is not a

concern in most applications, but can cause clipping if the

desired signal swing at the load is very close to the amplifier’s

swing limit. Such clipping would be most likely to occur in

pulse response applications where the frequency peaking is

manifested as an overshoot in the step response.

Parasitic capacitive loads greater than 2pF can begin to

degrade the performance of the OPA842. Long PC board

traces, unmatched cables, and connections to multiple de-

vices can easily cause this value to be exceeded. Always

consider this effect carefully, and add the recommended

series resistor as close as possible to the OPA842 output pin

(see Board Layout section).

DISTORTION PERFORMANCE

The OPA842 is capable of delivering an exceptionally low

distortion signal at high frequencies and low gains. The

distortion plots in the Typical Characteristics show the typical

distortion under a wide variety of conditions. Most of these

plots are limited to 100dB dynamic range. The OPA842’s

distortion does not rise above –100dBc until either the signal

level exceeds 0.5V and/or the fundamental frequency ex-

ceeds 500kHz. Distortion in the audio band is

≤ –120dBc.

Generally, until the fundamental signal reaches very high

frequencies or powers, the 2nd-harmonic will dominate the

distortion with a negligible 3rd-harmonic component. Focus-

ing then on the 2nd-harmonic, increasing the load imped-

ance improves distortion directly. Remember that the total

load includes the feedback network—in the noninverting

configuration this is the sum of RF + RG, whereas in the

inverting configuration this is just RF (see Figure 1). Increas-

ing the output voltage swing increases harmonic distortion

directly. Increasing the signal gain will also increase the 2nd-

harmonic distortion. Again, a 6dB increase in gain will in-

crease the 2nd- and 3rd-harmonic by 6dB even with a

constant output power and frequency. Finally, the distortion

increases as the fundamental frequency increases due to the

roll off in the loop gain with frequency. Conversely, the

distortion will improve going to lower frequencies down to the

dominant open-loop pole at approximately 600Hz. Starting

from the –100dBc 2nd-harmonic for 2Vp-p into 200

Ω, G = +2

distortion at 1MHz (from the Typical Characteristics), the

2nd-harmonic distortion at 20kHz should be approximately:

–100dB – 20log (1MHz/20kHz) = –134dBc

The OPA842 has an extremely low 3rd-order harmonic distor-

tion. This also gives an exceptionally good 2-tone, 3rd-order

intermodulation intercept, as shown in the Typical Character-

istics. This intercept curve is defined at the 50

Ω load when

driven through a 50

Ω-matching resistor to allow direct com-

parisons to RF MMIC devices. This network attenuates the

voltage swing from the output pin to the load by 6dB. If the

OPA842 drives directly into the input of a high-impedance

device, such as an ADC, this 6dB attenuation is not taken.

Under these conditions, the intercept will increase by a mini-

mum 6dBm. The intercept is used to predict the intermodulation

spurious for two closely spaced frequencies. If the two test

frequencies, f1 and f2, are specified in terms of average and

delta frequency, fO = (f1 + f2)/2 and ∆f = |f2 – f1|/2, the two 3rd-

order, close-in spurious tones will appear at fO ± (3 • ∆f). The

difference between the two equal test-tone power levels and

these intermodulation spurious power levels is given by

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