OPA690IDBVT datasheet(17/22 Pages) TI

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OPA690

SBOS223A

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levels, the 2nd-harmonic will dominate the distortion with a

negligible 3rd-harmonic component. Focusing then on the

2nd-harmonic, increasing the load impedance improves

distortion directly. Remember that the total load includes

the feedback network; in the noninverting configuration (see

Figure 1) this is sum of RF + RG, while in the inverting

configuration, it is just RF. Also, providing an additional

supply decoupling capacitor (0.1

µF) between the supply pins

(for bipolar operation) improves the 2nd-order distortion

slightly (3dB to 6dB).

In most op amps, increasing the output voltage swing in-

creases harmonic distortion directly. The new output stage

used in the OPA690 actually holds the difference between

fundamental power and the 2nd- and 3rd-harmonic powers

relatively constant with increasing output power until very

large output swings are required (> 4Vp-p). This also shows

up in the 2-tone, 3rd-order intermodulation spurious (IM3)

response curves. The 3rd-order spurious levels are moder-

ately low at low output power levels. The output stage

continues to hold them low even as the fundamental power

reaches very high levels. As the Typical Characteristics

show, the spurious intermodulation powers do not increase

as predicted by a traditional intercept model. As the funda-

mental power level increases, the dynamic range does not

decrease significantly. For 2 tones centered at 20MHz, with

10dBm/tone into a matched 50

Ω load (i.e., 2Vp-p for each

tone at the load, which requires 8Vp-p for the overall 2-tone

envelope at the output pin), the Typical Characteristics show

47dBc difference between the test tone powers and the 3rd-

order intermodulation spurious powers. This performance

improves further when operating at lower frequencies.

NOISE PERFORMANCE

High slew rate, unity-gain stable, voltage feedback op amps

usually achieve their slew rate at the expense of a higher

input noise voltage. The 5.5nV/

√Hz input voltage noise for

the OPA690 is, however, much lower than comparable

amplifiers. The input-referred voltage noise, and the two

input-referred current noise terms, combine to give low

output noise under a wide variety of operating conditions.

Figure 11 shows the op amp noise analysis model with all the noise

terms included. In this model, all noise terms are taken to be noise

voltage or current density terms in either nV/

√Hz or pA/√Hz.

The total output spot noise voltage can be computed as the

square root of the sum of all squared output noise voltage

contributors. Equation 1 shows the general form for the

output noise voltage using the terms shown in Figure 11.

(1)

kTR

I R

kTR NG

BN SS

BI F

(

) +









(

) +

Dividing this expression by the noise gain (NG = (1+RF/RG))

will give the equivalent input-referred spot noise voltage at the

noninverting input, as shown in Equation 2.

(2)

kTR

NNI

BN SS

BI F

(

) ++ 







Evaluating these two equations for the OPA690 circuit and

component values (see Figure 1) will give a total output spot

noise voltage of 12.3nV/

√Hz and a total equivalent input spot

noise voltage of 6.1nV/

√Hz. This is including the noise added

by the bias current cancellation resistor (175

Ω) on the

noninverting input. This total input-referred spot noise volt-

age is only slightly higher than the 5.5nV/

√Hz specification

for the op amp voltage noise alone. This will be the case as

long as the impedances appearing at each op amp input are

limited to the previously recommend maximum value of

300

Ω. Keeping both (R

F || RG) and the noninverting input

source impedance less than 300

Ω will satisfy both noise and

frequency response flatness considerations. Since the resis-

tor-induced noise is relatively negligible, additional capacitive

decoupling across the bias current cancellation resistor (RB)

for the inverting op amp configuration of Figure 8 is not

required.

DC ACCURACY AND OFFSET CONTROL

The balanced input stage of a wideband voltage feedback op

amp allows good output DC accuracy in a wide variety of

applications. The power-supply current trim for the OPA690

gives even tighter control than comparable products. Al-

though the high-speed input stage does require relatively

high input bias current (typically

±8µA at each input terminal),

the close matching between them may be used to reduce the

output DC error caused by this current. The total output offset

voltage may be considerably reduced by matching the DC

source resistances appearing at the two inputs. This reduces

the output DC error due to the input bias currents to the offset

current times the feedback resistor. Evaluating the configura-

tion of Figure 1, using worst-case +25

°C input offset voltage

and current specifications, gives a worst-case output offset

voltage equal to: – (NG = noninverting signal gain)

±(NG • V

OS(MAX)) ± (RF • IOS(MAX))

±(2 • 4mV) ± (402Ω • 1µA)

±8.4mV

FIGURE 11. Op Amp Noise Analysis Model.

4kT

OPA690

4kT = 1.6E –20J

at 290

°K

4kTR

√

4kTR

√

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